inherent in scaling make the design of radio frequency integrated circuits a. ebooks can be used on all reading devices; Immediate eBook download after. Radio Frequency Integrated Circuit Design For a listing of recent titles in the Artech House Microwave Library, turn. Radio Frequency Integrated Circuit Design (Artech House Microwave Library ( Hardcover)) [John Rogers, Calvin Plett] on ukraine-europe.info *FREE* shipping on.
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Radio-Frequency Integrated-Circuit Engineering addresses the theory, analysis and design of passive and active RFIC's using Si-based CMOS and Bi-CMOS. May 5, "The Design of CMOS Radio-Frequency Integrated Circuits" can you post the link to where download books is on the ukraine-europe.info website?. Advances in Analog and RF IC Design for Wireless Communication Systems - 1st Edition - ISBN eBook ISBN: Open - Buy once, receive and download all available eBook formats, including He is also member of the Steering Committee of the Radio Frequency Integrated Circuits Conference ( RFIC).
Radio frequency integrated circuit design Home Radio frequency integrated circuit design. ISBN x alk. Radio frequency integrated circuits—Design and construction. Very high speed integrated circuits. Plett, Calvin. R64 Radio circuits—Design and construction 2.
In my career, there has been a gap in comprehension between analog low-frequency designers and microwave designers. Often, similar issues were dealt with in two different languages. Although this book is more firmly based in lumped-element analog circuit design, it is nice to see that microwave knowledge is brought in where necessary.
Too many analog circuit books in the past have concentrated first on the circuit side rather than on basic theory behind their application in communications.
The circuits usually used have evolved through experience, without a satisfying intellectual theme in describing them. Why a given circuit works best can be subtle, and often these circuits are chosen only through experience. For this reason, I am happy that the book begins first with topics that require an intellectual approach—noise, linearity and filtering, and technology issues.
I am particularly happy with how linearity is introduced power series. In the rest of the book it is then shown, with specific circuits and numerical examples, how linearity and noise issues arise.
Concentration is on bipolar circuits, not metal oxide semiconductors MOS. Bipolar still has many advantages at high frequency. The depth with which design issues are addressed would not be possible if similar MOS coverage was attempted. However, there might be room for a similar book, which concentrates on MOS. In this book there is a lot of detailed academic exploration of some important high-frequency RF bipolar ICs. One might ask if this is important in design for application, and the answer is yes.
To understand why, one must appreciate the central role of analog circuit simulators in the design of such circuits. At the beginning of my career around — discrete circuits were large enough that good circuit topologies could be picked out by breadboarding with the actual parts themselves.
This worked fairly well with some analog circuits at audio frequencies, but failed completely in the progression to integrated circuits. In high-speed IC design nowadays, the computer-based circuit simulator is crucial.
Such simulation is important at four levels. The first level is the use of simplified models of the circuit elements idealized transistors, capacitors, and inductors. The use of such models allows one to pick out good topologies and eliminate bad ones. This is not done well with just paper analysis because it will miss key factors, such as the complexities of the transistor, particularly nonlinearity and bias and signal interaction effects.
Exploration of topologies with the aid of a circuit simulator is necessary. The simulator is useful for quick iteration of proposed circuits, with simplified models to show any fundamental problems with a proposed circuit. This brings out the influence of model parameters on circuit performance.
This first level of simulation may be avoided if the best topology, known through experience, is picked at the start. The second level of simulation is where the models are representative of the type of fabrication technology being used. However, we do not yet use specific numbers from the specific fabrication process and make an educated approximation to likely parasitic capacitances.
Simulation at this level can be used to home in on good values for circuit parameters for a given topology before the final fabrication process is available. Before the simulation begins, detailed preliminary analysis at the level of this book is possible, and many parameters can be wisely chosen before simulation begins, greatly shortening the design process and the required number of iterations.
Thus, the analysis should focus on topics that arise, given a typical fabrication process. I believe this has been done well here, and the authors, through scholarly work and real design experience, have chosen key circuits and topics.
The third level of design is where a link with a proprietary industrial process has been made, and good simulator models are supplied for the process. The circuit is laid out in the proprietary process and simulation is done, including Foreword xvii estimates of parasitic capacitances from interconnections and detailed models of the elements used. The incorporation of the proprietary models in the simulation of the circuit is necessary because when the IC is laid out in the actual process, fabrication of the result must be successful to the highest possible degree.
This is because fabrication and testing is extremely expensive, and any failure can result in the necessity to change the design, requiring further fabrication and retesting, causing delay in getting the product to market. The fourth design level is the comparison of the circuit behavior predicted from simulation with that of measurements of the actual circuit.
Discrepancies must be explained. These may be from design errors or from inadequacies in the models, which are uncovered by the experimental result. These model inadequacies, when corrected, may result in further simulation, which causes the circuit design and layout to be refined with further fabrication.
This discussion has served to bring attention to the central role that computer simulation has in the design of integrated RF circuits, and the accompanying importance of circuit analysis such as presented in this book. Such detailed analysis may save money by facilitating the early success of applications. This book can be beneficial to designers, or by those less focused on specific design, for recognizing key constraints in the area, with faith justified, I believe, that the book is a correct picture of the reality of high-speed RF communications circuit design.
Miles A. We decided that we would organize some of the research we and many others had been doing and turn it into a manuscript that would serve as a comprehensive text for engineers interested in learning about radio frequency integrated circuits RFIC. We have focused mainly on bipolar technology in the text, but since many techniques in RFICs are independent of technology, we hope that designers working with other technologies will also find much of the text useful.
We have tried very hard to identify and exterminate bugs and errors from the text. Undoubtedly there are still many remaining, so we ask you, the reader, for your understanding.
Please feel free to contact us with your comments. We hope that these pages add to your understanding of the subject. Nobody undertakes a project like this without support on a number of levels, and there are many people that we need to thank. Professors Miles Copeland and Garry Tarr provided technical guidance and editing. David, we have tried to add some of your wisdom to these pages. Thanks also go to Dave Rahn and Steve Kovacic, who have both contributed to our research efforts in a variety of ways.
We would like to thank Sandi Plett who tirelessly edited chapters, provided formatting, and helped beat the word processor into submission. She did more than anybody except the authors to make this project happen.
We would also like to thank a number of graduate students, alumni, and colleagues who have helped us with our understanding of RFICs over the years. Technologies are constantly being improved, and as they are, circuits formerly implemented as discrete solutions can now be integrated onto a single chip.
In addition to widely used applications such as cordless phones and cell phones, new applications continue to emerge. Thus, the market is expanding, and with each new application there are unique challenges for the designers to overcome. As a result, the field of RFIC design should have an abundance of products to keep designers entertained for years to come.
This huge increase in interest in radio frequency RF communications has resulted in an effort to provide components and complete systems on an integrated circuit IC. In academia, there has been much research aimed at putting a complete radio on one chip.
Since complementary metal oxide semiconductor CMOS is required for the digital signal processing DSP in the back end, much of this effort has been devoted to designing radios using CMOS technologies [1—3]. However, bipolar design continues to be the industry standard because it is a more developed technology and, in many cases, is better modeled. Major research is being done in this area as well. CMOS traditionally had the advantage of lower production cost, but as technology dimensions become 1 2 Radio Frequency Integrated Circuit Design smaller, this is becoming less true.
Which will win? Who is to say? Ultimately, both will probably be replaced by radically different technologies. In any case, as long as people want to communicate, engineers will still be building radios. Contrary to popular belief, most of the design concepts in RFIC design are applicable regardless of what technology is used to implement them.
The objective of a radio is to transmit or receive a signal between source and destination with acceptable quality and without incurring a high cost. From a more technical point of view, quality is often measured in terms of bit error rate, and acceptable quality might be to experience less than one error in every million bits. Cost can be seen as the price of the communications equipment or the need to replace or recharge batteries.
Low cost implies simple circuits to minimize circuit area, but also low power dissipation to maximize battery life.
The most fundamental difference between low-frequency analog and microwave design is that in microwave design, transmission line concepts are important, while in low-frequency analog design, they are not. This will have implications for the choice of impedance levels, as well as how signal size, noise, and distortion are described.
On-chip dimensions are small, so even at RF frequencies 0. However, at the chip boundaries, or when traversing a significant fraction of a wavelength on chip, transmission line theory becomes very important. Thus, on chip we can usually make use of analog design concepts, although, in practice, microwave design concepts are often used.
At the chip interfaces with the outside world, we must treat it like a microwave circuit. For example, an operational amplifier can be used as a buffer because its high input impedance does not affect the circuit to which it is connected, and its low output impedance Introduction to Communications Circuits 3 can drive a measurement device efficiently.
The freedom to choose arbitrary impedance levels provides advantages in that circuits can drive or be driven by an impedance that best suits them. On the other hand, if circuits are connected using transmission lines, then these circuits are usually designed to have an input and output impedance that match the characteristic impedance of the transmission line. In microwave circuits, power is usually used to describe signals, noise, or distortion with the typical unit of measure being decibels above 1 milliwatt dBm.
However, in analog circuits, since infinite or zero impedance is allowed, power levels are meaningless, so voltages and current are usually chosen to describe the signal levels.
Voltage and current are expressed as peak, peak-to-peak, or root-mean-square rms. Power in dBm, P dBm , can be related to the power in watts, Pwatt , as shown in 1. Note also that v rms can be related to the peak voltage v pp by Table 1. Noise is usually represented as noise density per hertz of bandwidth. In analog circuits, noise is specified as squared volts per hertz, or volts per square root of hertz. In both analog and microwave circuits, an effect of nonlinearity is the appearance of harmonic distortion or intermodulation distortion, often at new frequencies.
In low-frequency analog circuits, this is often described by the ratio of the distortion components compared to the fundamental components. In microwave circuits, the tendency is to describe distortion by gain compression power level where the gain is reduced due to nonlinearity or third-order intercept point IP3.
Noise and linearity are discussed in detail in Chapter 2. A summary of low-frequency analog and microwave design is shown in Table 1. Many aspects of this transceiver are common to all transceivers.
Table 1. This transceiver has a transmit side Tx and a receive side Rx , which are connected to the antenna through a duplexer that can be realized as a switch or a filter, depending on the communications standard being followed.
The input preselection filter takes the broad spectrum of signals coming from the antenna and removes the signals not in the band of interest. This may be required to prevent overloading of the low-noise amplifier LNA by out-ofband signals. The LNA amplifies the input signal without adding much noise. The input signal can be very weak, so the first thing to do is strengthen the signal without corrupting it.
As a result, noise added in later stages will be of less importance. The image filter that follows the LNA removes out-of-band signals and noise which will be discussed in detail in Chapter 2 before the signal enters the mixer.
Editions of The Design of CMOS Radio-Frequency Integrated Circuits by Thomas H. Lee
The mixer translates the input RF signal down to the intermediate frequency, since filtering, as well as circuit design, becomes much easier at lower frequencies for a multitude of reasons. The other input to the mixer is the local oscillator LO signal provided by a voltage-controlled oscillator inside a frequency synthesizer.
The desired output of the mixer will be the difference between the LO frequency and the RF frequency. At the input of the radio there may be many different channels or frequency bands.
The LO frequency is adjusted so that the desired RF channel or frequency band is mixed down to the same intermediate frequency IF in all cases. The IF stage then provides channel filtering at this one frequency to remove the unwanted channels. The IF stage provides further amplification and automatic gain control AGC to bring the signal to a specific amplitude level before the signal is passed on to the back end of the receiver. It will ultimately be converted into bits most modern communications systems use digital modulation schemes that could represent, for example, voice, video, or data through the use of an analog-to-digital converter.
In the IF stage, there may be some filtering to remove unwanted signals generated by the baseband, and the signal may or may not be converted into an analog waveform before it is modulated onto the IF carrier.
A mixer converts the modulated signal and IF carrier up to the desired RF frequency. A frequency synthesizer provides the other mixer input. Since the RF carrier and associated modulated data may have to be transmitted over large distances through lossy media e. Typically, the power level is increased from the milliwatt range to a level in the range of hundreds of milliwatts to watts, depending on the particular application.
A lowpass filter after the PA removes any harmonics produced by the PA to prevent them from also being transmitted.
Components are designed with the main concerns being frequency response, gain, stability, noise, distortion nonlinearity , impedance matching, and power dissipation. Dealing with design constraints is what keeps the RFIC designer employed. The focus of this book will be how to design and build the major circuit blocks that make up the RF portion of a radio using an IC technology.
To that end, block level performance specifications are described in Chapter 2.
Radio Frequency Integrated Circuits and Technologies
A brief overview of IC technologies and transistor performance is given in Chapter 3. Various methods of matching impedances, which are very important at chip boundaries and for some interconnections of circuits on-chip, will be discussed in Chapter 4. The realization and limitations of passive circuit components in an IC technology will be discussed in Chapter 5. Chapters 6 through 10 will be devoted to individual circuit blocks such as LNAs, mixers, voltage-controlled oscillators VCOs , filters, and power amplifiers.
However, the design of complete synthesizers is beyond the scope of this book. The interested reader is referred to [8—10]. References  Lee, T. Cambridge University Press, Prentice Hall, Introduction to Communications Circuits 7  Crols, J. Kluwer Academic Publishers, Artech House, Theory and Design, New York: IEEE Press, Nonidealities we will consider include noise and nonlinearity. We will also consider the effect of filtering. An ideal circuit, such as an amplifier, produces a perfect copy of the input signal at the output.
In a real circuit, the amplifier will introduce both noise and distortion to that waveform. Noise, which is present in all resistors and active devices, limits the minimum detectable signal in a radio. At the other amplitude extreme, nonlinearities in the circuit blocks will cause the output signal to become distorted, limiting the maximum signal amplitude. At the system level, specifications for linearity and noise as well as many other parameters must be determined before the circuit can be designed.
In this chapter, before we look at circuit details, we will look at some of these system issues in more detail. In order to design radio frequency integrated circuits with realistic specifications, we need to understand the impact of noise on minimum detectable signals and the effect of nonlinearity on distortion.
Knowledge of noise floors and distortion will be used to understand the requirements for circuit parameters. In addition to the desired signal, the receiver is also picking up noise from the rest of the universe. This thermal energy moves atoms and electrons around in a random way, leading to random currents in circuits, which are also noise. Noise can also come from man-made sources such as microwave ovens, cell phones, pagers, and radio antennas.
Circuit designers are mostly concerned with how much noise is being added by the circuits in the transceiver. At the input to the receiver, there will be some noise power present that defines the noise floor. The minimum detectable signal must be higher than the noise floor by some signal-to-noise ratio SNR to detect signals reliably and to compensate for additional noise added by circuitry.
These concepts will be described in the following sections. We note that to find the total noise due to a number of sources, the relationship of the sources with each other has to be considered. The most common assumption is that all noise sources are random and have no relationship with each other, so they are said to be uncorrelated. In such a case, noise power is added instead of noise voltage.
Similarly, if noise at different frequencies is uncorrelated, noise power is added. We note that signals, like noise, can also be uncorrelated, such as signals at different unrelated frequencies. In such a case, one finds the total output signal by adding the powers. On the other hand, if two sources are correlated, the voltages can be added.
As an example, correlated noise is seen at the outputs of two separate paths that have the same origin. Noise in resistors is generated by thermal energy causing random electron motion [1—3]. Noise power spectral density is expressed using volts squared per hertz power spectral density. In order to find out how much power a resistor produces in a finite bandwidth, simply multiply 2.
This can also be written equivalently as a noise current rather than a noise voltage: The model for noise in a resistor is shown in Figure 2. Note that kT is in watts per hertz, which is a power density. However, if the antenna were pointed at the sky, the equivalent noise temperature would be much lower, more typically 50K . For any receiver required to receive a given signal bandwidth, the minimum detectable signal can now be determined. As can be seen from 2.
The minimum detectable signal in a receiver is also referred to as the receiver sensitivity.
However, the SNR required to detect bits reliably e. The actual required SNR depends on a variety of factors, such as bit rate, energy per bit, IF filter bandwidth, detection method e. Such calculations are the topics for a digital communications course [6, 7] and will not be discussed further here. It should be noted that for data transmission, lower BER is often required e.
Thus, for reliable detection, the previously calculated minimum detectable signal level must be modified to include the noise from the active circuitry. Noise from the electronics is described by noise factor F, which is a measure of how much the signal-to-noise ratio is degraded through the system. We derive the following equation for noise factor: If N o source is the noise at the output originating at the source, and N o added is the noise at the output added by the electronic circuitry, then we can write: In other words, an electronic system that adds no noise has a noise figure of 0 dB.
In the receiver chain, for components with loss such as switches and filters , the noise figure is equal to the attenuation of the signal. This is explained by noting that output noise is approximately equal to input noise, but signal is attenuated by 3 dB.
Thus, there has been a degradation of SNR by 3 dB. For the purposes of developing 2. It will be shown in later chapters how to take a practical amplifier and make it fit this model. In this model, all noise sources in the circuit are lumped into a series noise voltage source v n and a parallel current noise source i n placed in front of a noiseless transfer function.
Figure 2. In general, they will not be correlated with each other, but rather the current i n will be partially correlated with v n and partially uncorrelated. We can expand both current and voltage into these two explicit parts: Equation 2. Differentiating with respect to G s and B s and setting the derivative to zero yields the following two conditions for minimum noise G opt and B opt after several pages of math: For this reason, we typically design systems with a low-noise amplifier at the front of the system.
We note that the noise figure of each block is typically determined for the case in which a standard input source e. The above formula can also be used to derive an equivalent model of each block as shown in Figure 2. Example 2. Determine how much noise voltage per unit bandwidth is present at the output.
Then, for any R L , what is the maximum noise power that this source can deliver to any load? Also find the noise factor, assuming that R L does not contribute to noise factor, and compare to the case where R L does contribute to noise factor.
Then the complete available power from the source is delivered to the load. Therefore, for a resistively matched circuit, the noise figure is 3 dB. Note that the output noise voltage is 0. Determine the noise at the output of the circuit due to all resistors and then determine the circuit noise figure and signal-to-noise ratio assuming a 1-MHz bandwidth and the input is a 1-V sine wave.
Solution In this example, at v x the noise is still due to only R S and R 2. As before, the noise at this point is 0. The signal at this point is 0. At the output, the signal and noise from the input sources, as well as the noise from the two output resistors, all see a voltage divider.
Thus, one can calculate the individual components. Each resistor at the input provides 4. Thus, as explained earlier, after a gain stage, noise is less important. Solution As before, the output noise due to the resistors is as follows: This circuit is unmatched at the input.
This example illustrates that a mismatched circuit may have better noise performance than a matched one. However, this assumes that it is possible to build a voltage amplifier that requires little power at the input. This may be possible on an IC. However, if transmission lines are included, power transfer will suffer.
A matching circuit may need to be added. The system consists of a filter with 3-dB loss, followed by a switch with 1-dB loss, an LNA, and a mixer. Also assume that the system bandwidth is kHz. Solution Since the bandwidth of the system has been given as kHz, the noise floor of the system can be determined: Furthermore, the noise figure of 12 dB corresponds to a noise factor of Note that if the mixer also has gain, then possibly the noise due to the IF stage may be ignored.
In a real system this would have to be checked, but here we will ignore noise in the IF stage. Since it was stated that the system requires an SNR of 7 dB, the sensitivity of the system can now be determined: If this is not adequate for a given application, then a number of things can be done to improve this: A smaller bandwidth could be used.
This is usually fixed by IF requirements. The loss in the preselect filter or switch could be reduced. For example, the LNA could be placed in front of one or both of these components. The noise figure of the LNA could be improved. A lower NF in the mixer would also improve the system NF. However, in any real device the transfer function is usually a lot more complicated. This can be due to active or passive devices in the circuit or the signal swing being limited by the power supply rails.
Unavoidably, the gain curve for any component is never a perfectly straight line, as illustrated in Figure 2. The resulting waveforms can appear as shown in Figure 2. For amplifier saturation, typically the top and bottom portions of the waveform are clipped equally, as shown in Figure 2.
However, if the circuit is not biased between the two clipping levels, then clipping can be nonsymmetrical as shown in Figure 2. Symmetrical saturation as shown in Figure 2. In another example, an exponential nonlinearity as shown in Figure 2. Real circuits will have more complex power series expansions. One common way of characterizing the linearity of a circuit is called the two-tone test. In this test, an input consisting of two sine waves is applied to the circuit.
For instance, the X 12 term has a zero frequency dc component and another at the second harmonic of the input: The mixing components will appear at the sum and difference frequencies of the two input signals. Note also that second-order terms cause an additional dc term to appear. The third-order terms can be expanded as follows: Expansion of both the HD3 and IM3 terms shows output signals appearing at the input frequencies.
The effect is that third-order nonlinearity can change the gain, which is seen as gain compression. This is summarized in Table 2. These two tones are usually referred to as third-order intermodulation terms IM3 products.
Determine all output frequency components, assuming distortion components up to the third order. Solution Table 2. It is apparent that harmonics can be filtered out easily, while the thirdorder intermodulation terms, being close to the desired tones, may be difficult to filter.
From the plot, the third-order intercept point IP3 is determined. From Table 2. A theoretical voltage at which these two tones will be equal can be defined: The input power at this point is called the input third-order intercept point IIP3. Issues in RFIC Design, Noise, Linearity, and Filtering 29 Of course, the third-order intercept point cannot actually be measured directly, since by the time the amplifier reached this point, it would be heavily overloaded.
Therefore, it is useful to describe a quick way to extrapolate it at a given power level. Assume that a device with power gain G has been measured to have an output power of P 1 at the fundamental frequency and a power of P 3 at the IM3 frequency for a given input power of P i , as illustrated in Figure 2. Now, on a log plot for example, when power is in dBm of P 3 and P 1 versus P i , the IM3 terms have a slope of 3 and the fundamental terms have a slope of 1. Which one is used depends largely on which is more important in the system of interest; for example, second-order distortion is particularly important in direct downconversion receivers.
The theoretical voltage at which the IM2 term will be equal to the fundamental term given in 2. This point is more directly measurable than IP3 and requires only one tone rather than two although any number of tones can be used. The 1-dB compression point is simply the power level, specified at either the input or the output, where the output power is 1 dB less than it would have been in an ideally linear device.
It is also marked in Figure 2. Solving 2. Since we now have expressions for both these values, we can find a relationship between these two points. Taking the ratio of 2. In the case of the 1-dB compression point with two tones applied, the ratio is larger. Thus, one can estimate that for a single tone, the compression point is about 10 dB below the intercept point, while for two tones, the 1-dB compression point is close to 15 dB below the intercept point.
The difference between these two numbers is just the factor of three 4. Note that this analysis is valid for third-order nonlinearity. For stronger nonlinearity i. Nevertheless, the above is a good estimate of performance. One is at a frequency of 2. At the output, four tones are observed at 1. Determine the IIP3 and 1-dB compression point for this amplifier. Solution The tones at 1. We can use 2. Two other measures of linearity that are common in wideband systems handling many signals simultaneously are called composite tripleorder beat CTB and composite second-order beat CSO [11, 12].
In these tests of linearity, N signals of voltage v i are applied to the circuit equally spaced in frequency, as shown in Figure 2. Note here that, as an example, the tones are spaced 6 MHz apart this is the spacing for a cable television system for which this is a popular way to characterize linearity. Note also that the tones are never placed at a frequency that is an exact multiple of the spacing in this case, 6 MHz.
This is done so that third-order terms and second-order terms fall at different frequencies. This will be clarified shortly. If we take three of these signals, then the third-order nonlinearity gets a little more complicated than before: In fact, there will be many more triple-beat TB products than IM3 products. Thus, these terms become more important in a wide-band system.
It can be shown that the maximum number of terms will fall on the tone at the middle of the band. We also note here that if the signal power is backed off from the IP3 power by some amount, the power in the IP3 tones will be backed off three times as much calculated on a logarithmic scale.
In this case, the signals fall at frequencies either above or below the carriers rather than right on top of them, as in the case of the triple-beat terms, provided that the carriers are not some even multiple of the channel spacing. For example, in Figure 2. This is 1. All the sum terms will fall 1.
Thus, the second-order and third-order terms can be measured separately. The number of terms that fall next to any given carrier will vary. For the case of the difference frequency second-order beats, there are more of these at lower frequencies, and the maximum number will be next to the lowest frequency carrier.
Noise determines how small a signal a receiver can handle, while linearity determines how large a signal a receiver can handle. This is illustrated in Figure 2. Determine the dynamic range of this receiver. Solution The overall receiver has a gain of 19 dB. The minimum detectable signal from Example 2.
Therefore, the mixer dominates the IIP3 for the receiver. The 1-dB compression point will be 9. What does this do to the dynamic range of the receiver? Solution This system is the same as the last one except that now the bandwidth is 80 MHz. In order to get this back to the value in the previous system, we would need to increase the linearity of the receiver by As we will see in future chapters, this would be no easy task.
The system bandwidth is set by filters, so it becomes necessary to discuss some of the filtering issues. There are additional reasons for needing filtering. The receiver must be able to maintain operation and to detect the desired signal in the presence of other signals often referred to as blocking signals. These other signals could be of large amplitude and could be close by in frequency.
Such signals must be removed by filters, so a very general discussion of filters is in order. Actual monolithic filter circuits will be discussed in a later chapter. A receiver in which the signal is taken directly to base band is called a homodyne or directconversion receiver.
Although simpler than a receiver that takes the signal to some IF first called a superheterodyne receiver , direct-conversion receivers suffer from numerous problems, including dc offsets, because much of the information is close to dc and also because of LO self-mixing .
An alternative to the image filter is to use an image reject mixer, which will be discussed in detail in Chapter 7. The image filter is required to suppress the unwanted image frequency, which is located a distance of two IFs away from the desired radio frequency .
Thomas Lee The Design of CMOS Radio-Frequency Ebook
Also, the image filter must prevent noise at the image frequency from mixing down to the IF and increasing the noise figure. A superheterodyne receiver takes the desired RF input signal and mixes it with some reference signal to extract the difference frequency, as shown in Figure 2. The problem is that a signal on the other side of the LO at the same distance from the LO will also mix down Figure 2.
Thus, before mixing can take place, this unwanted image frequency must be removed. Typically, this is done with a filter that attenuates the image.
Thus, another important specification in a receiver is how much image rejection it has. Image rejection is defined as the ratio of the gain of the desired signal through the receiver G sig to the gain of the image signal through the receiver G im. The following equation can be used for this calculation: The first IF is at 70 MHz.
Solution The frequency spectrum is shown in Figure 2. This is known as the image frequency. An image reject filter is required to prevent any image signals from entering the mixer.
Shuang-Hua Yang. Khaled Elleithy. Physical Layer Multi-Core Prototyping. Maxime Pelcat. Wenhua Chen. Qing Duan. Jagannath Malik.
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Originally Posted by mckinson. Thank you very much. You can use winrar compress the file and divide it into several parts. Then you can upload to this site. You can also upload to MCU. Anyone can upload? I am not able to view your link to 21rf It is in Chinese, so I just suppose that. Design of CMOS radio-frequency integrated circuits solutions 0.
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